Space-time coding digital transmission systems and methods

ABSTRACT

The invention concerns digital signal transmission. In particular, it concerns high speed transmission using layered space-time encoding architecture adapted to all types of propagation channels.  
     The invention therefore proposes a digital signal transmission system comprising:  
     a space-time encoder ( 1 ) receiving a flow of data to be transmitted d[i], formatting this data d[i] as symbol vectors  v [k] of dimension P (P&gt;1) and generating said symbol vectors  v [k], and  
     modulator-transmitters { 2   p } (1≦p≦P) , each receiving one component of the symbol vector  m [k] output from the space-time encoder ( 1 ), applying the constellation of a predetermined modulation to said symbol m p [k], and converting the symbol obtained a p [k] into a signal s p (t) presenting time diversity transmitted on said antenna ( 24   p ) connected to said transmitter ( 2   p ).  
     To demodulate in parallel the Q signals of the space-time observation  
           y   _          [   k   ]       =         ∑     j   =   0       J   -   1                           H   y          (   jTs   )       ·       a   _          [     k   -   j     ]           +         b   _     y          [   k   ]                       
 
     where  a [k] is the symbol vector transmitted at instant t=kTs+i, H y (t) the transfer function taking into account at least the transmission-reception, modulation, channel filters and the transmission-reception antenna gains and  b   y (t) the noise, the invention proposes a two dimensional suitable estimator-demodulator.

[0001] The invention concerns digital signal transmission. In particular, it concerns high speed transmission using layered space-time encoding architecture adapted to all types of propagation channel.

[0002] Traditionally, digital signal transmission is carried out using a system formed from a single transmission antenna and a single reception antenna. The objective is to improve the transmission speed, i.e. to transmit data bits (or symbols) between a transmission system and a reception system with a very high data rate. To do this, Bell Labs proposed the BLAST (Bell Labs Layered Space-Time) architecture that uses in transmission a system of P>1 antennas transmitting independent symbols and in reception a system of N≧P antennas.

[0003]FIG. 1 shows a transmission-reception system using BLAST architecture. The data d[i] to be transmitted is encoded as symbol vectors a[k]=[a₁[k] . . . a_(p)[k])^(T) by the space-time encoder 1. The symbol a_(p)[k] is the k^(th) symbol transmitted by the p^(th) transmitter 2 ^(p) (1≦p≦P).

[0004] The dimension of symbol vector a _(p)[k] is P corresponding to the number P of antennas in the transmission antenna network. These symbol vectors a _(p)[k] are processed then transmitted as signal vectors s[t] of dimension P by the P modulator-transmitters {2 ^(p)}_((1≦p≦P)) on its transmission antenna network {24^(p)}_((1≦p≦P)).

[0005] The signal model in the following expression used in the BLAST architecture is that of a signal with no time memory. In fact, the signal s[k] of symbols transmitted at instant k depends only on the symbols a[k] transmitted at the same instant by the P modulator-transmitters {2 ^(p)}_((1≦p≦P)). ${\underset{\_}{s}(k)} = {\begin{bmatrix} {s_{1}(k)} \\ \vdots \\ {s_{P}(k)} \end{bmatrix} = {{\left\lbrack {h_{1}\cdots \quad h_{P}} \right\rbrack \cdot \begin{bmatrix} {a_{1}(k)} \\ \vdots \\ {a_{P}(k)} \end{bmatrix}} = {{\underset{\_}{h}(k)} \cdot {\underset{\_}{a}(k)}}}}$

[0006] where h_(p) is the transmission filter of the p^(th) transmitter.

[0007] Under these conditions, the data rate can be increased by a factor P since P series of independent symbols are transmitted in parallel. The signals s(t) so transmitted follow M paths (M≧1) and are received by the N antennas of the reception antenna network. Receiver 3 generates the signal vector x(t), of dimension N, received by its antenna network associated with the space-time decoder 4 that can estimate, demodulate and decode the symbols a(k) transmitted, from which it deduces an estimation of the data d[i] transmitted.

[0008] Assuming that the transmitted signal is a linear modulation and that this signal is received at the symbol rate, the input-output relation between the transmitters and the receivers is as follows: ${\underset{\_}{x}(k)} = {\begin{bmatrix} {x_{1}(k)} \\ \vdots \\ {x_{N}(k)} \end{bmatrix} = {{{H \cdot \begin{bmatrix} {a_{1}(k)} \\ \vdots \\ {a_{P}(k)} \end{bmatrix}} + {\underset{\_}{b}(k)}} = {{H \cdot {\underset{\_}{a}(k)}} + {\underset{\_}{b}(k)}}}}$

[0009] where a[k] is a vector including the symbols transmitted in parallel, H the transfer function between transmission and reception, x[k] a vector including the received signals and b[k] the additive noise.

[0010] The space-time decoder 4 includes a signal processing system that can estimate the symbols a_(p)[k]. To estimate the p^(th) symbol a_(p)[k] using the above equation, the following spatial filtering is carried out: â_(p)(k)=w _(p) ^(t)·x(k).

[0011] To estimate the weighting vector w _(p), the article “An architecture for realizing very high data rates over the rich-scattering wireless channel” by Wolniansky, Foschini, Golden and Valenzuela, Proc. ISSE-98, Pisa, Italy, 29 Sep. 1998 summarizes two of these traditional linear estimation detection techniques using the BLAST algorithm estimating this filter. Consequently, by putting H=[h(1) . . . h(p)], the following two techniques can be carried out:

[0012] the jammer cancellation technique: w _(p l is the solution to the equation system w) _(p) ^(t) h(i)=δ_(pi) for 1≦i≦P. The sign δ_(pi) is the Kronecker symbol satisfying δ_(pi)=1 for p=i and δ_(pi)=0 for p≠1.

[0013] the technique maximizing the signal to noise ratio and jammer: the spatial filter must maximize the energy of {overscore (a)}_(p)[k] given that the useful symbol is then a_(p)[k] and the jammer symbols are the other symbols a_(i)[k] such that i≠p.

[0014] After estimating â_(p)[k], the state of the symbol â_(p)[k] is detected and the symbol {overscore (a)}[k] is deduced. With BPSK (Bi-Phase Shift Keying modulation) the decision is made between the phases 0 or π of the estimated symbol â[k]. Once decided, the decoded symbol ã[k] is demodulated, the demodulated symbol is equal to 1 or −1.

[0015] This linear technique is a readaptation to the BLAST architecture of the MMSE (Minimum Mean Square Error) type linear equalization technique. In this case, a spatial filter, therefore always one-dimensional, is used instead of the time filter to make the estimation.

[0016] To improve the linear technique, a DFE (Decision Feed-back Equalization) type algorithm is used to carry out the spatial filtering non-linearly. Under these conditions, the components of the symbol vector a[k] are estimated one by one, the estimated and detected symbol a_(p)[k] being removed from the spatial observation vector x[k] before estimating the next symbol a_(p+1)[k].

[0017] The company Bell Labs designed two techniques based on this principle. The first called V-BLAST is described in the article “An architecture for realizing very high data rates over the rich-scattering wireless channel” by Wolnianski, Foschini, Golden and Valenzuela, Proc. ISSE-98, Pisa, Italy, 29 Sep. 1998.

[0018] At each instant k, all components a_(p)[k] of the symbol vector a(k) are estimated and detected. By considering the time k as abscissa and the index p of the transmission sensor as ordinate, the estimation-detection is therefore carried out in the vertical direction, hence the name V-BLAST. By putting H=[h(1) . . . h(p)], the estimation-detection is carried out in the direction {p₁,p_(p)} such that h(p₁)^(t) h(p₁)> . . . >h(p_(p))^(t) h(p_(p)) . The V-BLAST estimation-detection algorithm is therefore carried out according to the following steps:

[0019] Initialization: i=1 and x ¹[k]=x[k],

[0020] In step i: Estimation and detection of the symbol a_(p)[k]:

â_(pi)(k)=w _(pi) ^(t)·x ^(i)(k)

ã_(pi)(k)

[0021] Cancellation of the symbol {overscore (a)}_(pi)[k] of the observations x[k]:

x ^(i+1)[k]=x ^(i)[k]−h(p_(i)) ã_(p)[k]

[0022] Stop: Move to next instant k=k+1, when i=P,

[0023] The second technique was the subject of two European patents EP 0 817 401 and EP 0 951 091. The non-linear estimation-detection algorithm described, the algorithm D-BLAST, only differs from the previous algorithm V-BLAST in that the direction of the estimation-detection of symbols {overscore (a)}_(pi)[k] is diagonal and no longer vertical.

[0024] The non-linear estimation-detection V-BLAST and D-BLAST can only be carried out under certain conditions. These conditions are as follows:

[0025] linear modulation without time memory,

[0026] demodulation on sampled signals at symbol rate,

[0027] transmission of synchronous independent symbols by the P modulator-transmitters,

[0028] number of receivers greater than or equal to the number of transmitters (N≧P),

[0029] network of transmission-reception antennas either non-colocalized or colocalized with a number of transmitters less than or equal to the number of paths (P≦M), given that a network of colocalized transmission-reception antennas is a network such that the dimension of the transmission antenna network and the dimension of the reception antenna network are much less than the distance between the transmission network and the reception network.

[0030] The relation between the transmitted symbols and the received symbols is therefore purely spatial.

[0031] This invention avoids or at least reduces these disadvantages, by proposing a transmission system of P modulator-transmitters transmitting symbols that can be estimated on reception with colocalized transmission and reception antenna networks irrespective of the transmission-reception and propagation conditions (modulation, disturbance, etc.).

[0032] A first objective is therefore to be able to also estimate the P series of symbols transmitted for a slightly disturbed propagation channel. The relation between the transmitted symbols and the received symbols is purely spatial and, for a slightly disturbed propagation channel, the spatial diversity is non-existent or virtually non-existent.

[0033] The invention therefore proposes a digital signal transmission system comprising:

[0034] a space-time encoder receiving a flow of data to be transmitted d[i], encoding this data d[i] as symbol vectors m[k] of dimension P (P>1) and generating said symbol vectors m[k], and

[0035] modulator-transmitters {2 ^(p)}_((1≦p≦P)), each receiving one component of the symbol vector m[k] output from the space-time encoder, applying the constellation of a predetermined modulation to said symbol m_(p)[k], and converting the symbol obtained a_(p)[k] into a signal s_(p)(t) transmitted on said antenna connected to said transmitter

[0036] wherein the transmitters are adapted to transmit signals s(t) with time diversity.

[0037] This transmission system operating, for example, via a digital signal transmission method comprising:

[0038] a space-time encoding step comprising at least the formatting as symbol vectors m[k] of dimension P (P>1) of the flow of data to be transmitted d[i], and

[0039] a modulator-transmission step comprising at least:

[0040] application in parallel of the constellation of a predetermined modulation to the P symbols m[k],

[0041] transmission in parallel of the P signals s(t) obtained from the constellated symbols a[k] from P spatially separate points, wherein the modulation-transmission step is adapted to transmit the signals s(t) with time diversity.

[0042] In order to estimate the P symbols so transmitted, the invention concerns an estimator-demodulator receiving in parallel N signals v(t) formed from L samples, wherein these signals v(t) represent a space-time observation since each of the N spatial components comprises L samples.

[0043] The estimator-demodulator previously described uses, for example, an estimation and demodulation method comprising a step of reception in parallel of N signals v(t) wherein the observation v(t) is space-time since each of the N spatial components comprises L samples.

[0044] The advantages and features of the invention will be clearer on reading the following description, given as an example, illustrated by the attached figures representing in:

[0045]FIG. 1, a transmission-reception system with BLAST type architecture according to the state of the art,

[0046]FIG. 2, an example of transmission system according to the invention,

[0047]FIGS. 3a, 3 b and 3 c some examples of filtering by the modulator-transmitters of the transmission system according to the invention,

[0048]FIG. 4, an example of a receiver according to the invention,

[0049]FIGS. 5a and 5 b some examples of estimation and decoding systems according to the invention.

[0050] In a transmission-reception system according to the invention, the useful data d[i] is formatted as a vector of dimension P by the device 11 in a space-time encoder 1, as shown on FIG. 2. The data vectors m[k] so obtained can then be encoded {12¹ . . . 12^(P)}. Apprenticeship sequences app known by the receiver in the device 13 are added to the symbol vectors c[k] o obtained. The symbols v[k] so obtained are then modulated by the modulation and transmission devices {2 ¹ . . . 2 ^(P)}. The devices {21 ¹ . . . 21 ^(P)} apply the chosen modulation constellation (for example the constellation-1, +1 with BPSK modulation) and generate the resulting symbol vector a[k].

[0051] Each symbol of the vector a[k] so obtained, the symbols a[k] representing the modulation states, can, using the device 22 ^(p) of the modulator-transmifter 2 ^(p), be formatted as a signal vector U _(p) ^(k)(t): ${{\underset{\_}{u}}_{p}^{K}\left( {t = {{kTs} + i}} \right)} = {\begin{bmatrix} {u_{p}(t)} \\ \vdots \\ {u_{p}\left( {t - K + 1} \right)} \end{bmatrix} = {{\begin{bmatrix} {\underset{-}{0}}_{i} \\ {a_{p}\lbrack k\rbrack} \\ {\underset{-}{0}}_{{Ts} - 1} \\ {a_{p}\left\lbrack {k - 1} \right\rbrack} \\ \vdots \end{bmatrix}\quad {for}\quad 0} \leq i < {Ts}}}$

[0052] with 0 _(T)=[0 . . . 0]^(T) and dim(0 _(T))=T×1 with T=i or T=Ts−1 where T_(s) is the symbol time. The realization of this vector U_(p) ^(k)(t) represents an oversampling of the symbols a_(p)[k] in order to satisfy Shannon's theorem. The vector U_(p) ^(k)(t) is then filtered by the formatting filter of device 22 ^(p). These filters {22 ¹ . . . 22 ^(P)} are the formatting filters of the chosen modulation (Gaussian filter, for example, with GMSK type modulation) and/or the transmission filter as such (wave formatting filter of type Nyquist, NRZ, etc.) and/or any other filter contained by the modulator-transmitters {2 ¹ . . . 2 ^(P)}. This device 22 ^(p) forms a filter whose continuous time function is h_(p)(t) (0≦t≦K, τ≧0): $\begin{matrix} {{s_{p}\left( {t - \tau} \right)} = {\sum\limits_{i = 0}^{K - 1}\quad {{h_{p}\left( {i - \tau} \right)}{u_{p}\left( {t - i} \right)}}}} \\ {= {\left\lbrack {{h_{p}\left( {- \tau} \right)}\quad \cdots \quad {h_{p}\left( {K - 1 - \tau} \right)}} \right\rbrack \cdot \begin{bmatrix} {u_{p}(t)} \\ \vdots \\ {u_{p}\left( {t - K + 1} \right)} \end{bmatrix}}} \\ {= {{{\underset{\_}{h}}_{p}(\tau)}^{T}{{\underset{\_}{u}}_{p}^{K}(t)}}} \end{matrix}$

[0053] The signal s_(p)(t) resulting from this filtering is transmitted by the p^(th) antenna 24 ^(p) of the transmission antenna network, after modulation with carrier frequency f₀ using device 23 ^(p). The signals r_(p)(t) modulated by a carrier frequency f₀ then give the transmission signals s_(p)(t) according to the relation:

s _(p)(t)=r _(p)(t)* exp(j2πf ₀ t),

[0054] The P modulator-transmitters {2 ^(p)}_((1≦p≦P)) then transmit signals related to independent symbols.

[0055] Device 13 used to add apprenticeship sequences can also be positioned before device 11, between device 11 and the encoders {12 ¹ . . . 12 ^(P)} or even before or after the modulation constellation application devices {21 ¹ . . . 21 ^(P)} or the filters {22 ¹ . . . 22 ^(P)}, etc.

[0056] The modulators {2 ¹ . . . 2 ^(P)} can be linear or linearizable, and with or without memory. For a linear modulator without memory, the signal s_(p)(t) depends only on the symbols a[k] at instant k. For modulation with time memory of dimension K, the signal s_(p)(t) also depends on vectors a[k−1] to a[k−K] (K≧1).

[0057] The filters {h_(p)(t)}_((1≦p≦P)) are all different from each other so that the receiver can also operate for a propagation channel with networks of colocalized antennas when the number of transmitters is greater than the number of paths (P≧M), especially for a single path propagation channel.

[0058]FIGS. 3a, 3 b and 3 c show examples of realization of these different filters {h_(p)(t)}_((1≦p≦P)) in order to meet this condition of time diversity of the P modulator-transmitters {2 ^(p}) _((1≦p≦P)).

[0059] This time diversity can be created in various ways:

[0060] by desynchronizing the signals transmitted by the P modulator-transmitters {2 ^(p)}_((1≦p≦P)),

[0061] by filtering with filters {22 ^(p)}_((1≦p≦P)) of different types: Nyquist, NRZ, etc. the symbols transmitted by the P modulator-transmitters,

[0062] by transmitting the signals s(t) transmitted by the P modulator-transmitters {2 ^(p)}(1≦p≦P) on different carrier frequencies{f_(p)}_((1≦p≦P)), spectrum overlap between the various transmitters being possible unlike with OFDM (Orthogonal Frequency Division Multiplexing),

[0063] etc.

[0064] On FIG. 3a, each filter h_(p)(t) comprises an element giving the type h of the filter and a delay element τ_(p) with τ₁ ≠τ₂. . . ≠τ_(p) such that h_(p)(t)=h(t−τ_(p)) for all values of p.

[0065] On FIG. 3b, the types h_(p) of the filters are all different from each other (h₁≠h₂≠ . . . h_(p)). Nyquist filters with roll-off α, NRZ filters, etc. can be used. $\begin{matrix} {{{{NRZ}:\quad {h_{p}(t)}} = {\Pi_{Ts}(t)}},{i.e.}} & {{{h_{p}(t)} = {{1\quad {if}\quad {t}} < {{Ts}/2}}}\quad} \\ {and} & {{h_{p}(t)} = {{0\quad {if}\quad {t}} < {{Ts}/2}}} \\ {{{Nyquist}\quad {with}\quad {roll}\text{-}{off}\quad {\alpha:\quad {h_{p}(t)}}} = {\frac{\cos \frac{\pi \quad \alpha \quad t}{Ts}}{1 - {4\frac{\alpha^{2}t^{2}}{{Ts}^{2}}}}\sin \quad {c\left( \frac{\pi \quad t}{Ts} \right)}}} & \quad \end{matrix}$

[0066] The filters h_(p) can, for example, all be Nyquist filters with different roll-off values α_(p).

[0067] On FIG. 3c, each filter h_(p)(t) comprises an element giving the type h of the filter and an element used to give a frequency shift to the signal r_(p)(t), with h_(p)(t)=h·exp(j2πf_(p)t), such that the frequencies are all different f₁≠f₂. . . f_(p).

[0068] We will consider the case of networks of colocalized transmission and reception antennas, the transmission antenna 24 ^(p) of the modulator-transmitter 2 ^(p) sends a signal s_(p)(t) which takes, for example, M paths as M plane waves of incidence θ_(m) ^(T) (1≦m≧M) that the N reception antennas of receiver 3 receive as M plane waves of incidence θhd m^(R) as shown on FIG. 1.

[0069] Under these conditions, the signals x(t) observed by receiver 3 of the receiver on FIG. 4 can be expressed as follows: ${\underset{\_}{x}(t)} = {{\sum\limits_{p = 1}^{P}\quad {\sum\limits_{m = 1}^{M}\quad {\rho_{m} \cdot {G_{p}\left( \theta_{m}^{T} \right)} \cdot {\underset{\_}{d}\left( \theta_{m}^{R} \right)} \cdot {s_{p}\left( {t - \tau_{m}} \right)}}}} + {\underset{\_}{b}(t)}}$

[0070] where τ_(m) and ρ_(m) are respectively the delay and the attenuation of the m^(th) path with respect to the direct path. The signal s_(p)(t) depends on the transmitted symbols a[k] contained in the vectors u _(p) ^(k)(t) according to the relations given in the description on FIG. 2. The signal x(t) can then be expressed, according to the symbol vectors u _(p) ^(k)(t) for 1≦p≦P: $\begin{matrix} {{\underset{\_}{x}(t)} = {{\sum\limits_{p = 1}^{P}\quad {\sum\limits_{m = 1}^{M}\quad {\rho_{m} \cdot {G_{p}\left( \theta_{m}^{T} \right)} \cdot {\underset{\_}{d}\left( \theta_{m}^{R} \right)} \cdot {{\underset{\_}{h}}_{p}\left( \tau_{m} \right)} \cdot {{\underset{\_}{u}}_{p}^{K}(t)}}}} + {\underset{\_}{b}(t)}}} \\ {= {{\sum\limits_{p = 1}^{P}\quad {H_{p} \cdot {{\underset{\_}{u}}_{p}^{K}(t)}}} + {\underset{\_}{b}(t)}}} \end{matrix}$

[0071] This last expression shows that the transfer functions H_(p) of the P modulator-transmitters {2 ^(p)}_((1≦p≦P)) differ in the filter of function h_(p)(τ_(m)) and the gain G_(p)(θ_(m) ^(T)) of the transmission antenna 24 ^(p).

[0072] The observation x(t) is transmitted by the various reception devices and filters {31 ^(n)}_((1≦n≦N)), comprising at least a carrier recovery device used to put the received signal in the baseband with one windower 32 $\begin{matrix} {{\underset{\_}{x}\left( {{kTs} + i} \right)} = {{\sum\limits_{j = 0}^{J - 1}\quad {\left\lbrack {{H_{1}\left( {i + {jTs}} \right)}\quad \cdots \quad {H_{p}\left( {i + {jTs}} \right)}} \right\rbrack \cdot \begin{bmatrix} {a_{1}\left\lbrack {k - j} \right\rbrack} \\ \vdots \\ {a_{P}\left\lbrack {k - j} \right\rbrack} \end{bmatrix}}} + {\underset{\_}{b}(t)}}} \\ {= {{\sum\limits_{j = 0}^{J - 1}\quad {{H\left( {i + {jTs}} \right)} \cdot {\underset{\_}{a}\left\lbrack {k - j} \right\rbrack}}} + {\underset{\_}{b}(t)}}} \end{matrix}$

[0073] where H_(p)(j) is the j^(th) column of matrix H_(p).

[0074] To better identify the vector a[k], in fact, device 32 windows the spatial observation x(t) so that a space-time observation y(t) is obtained. Given that the vectors x(kTs+i), with 0≦i>Ts, depend on the symbol vectors a[k] to a[k−J+1], the next vector v(t) is formed. $\begin{matrix} {{\underset{\_}{y}(t)} = \begin{bmatrix} {\underset{\_}{x}(t)} \\ {\underset{\_}{x}\left( {t - 1} \right)} \\ \vdots \\ {\underset{\_}{x}\left( {t + L - 1} \right)} \end{bmatrix}} \\ \begin{matrix} {{\underset{\_}{y}\left( {t = {{kTs} + i}} \right)} = \begin{bmatrix} {\underset{\_}{x}\left( {{kTs} = i} \right)} \\ \vdots \\ {\underset{\_}{x}\left( {{kTs} + i + L - 1} \right)} \end{bmatrix}} \\ {= {{\sum\limits_{j = 0}^{J - 1}\quad {\begin{bmatrix} {H\left( {{jTs} + i} \right)} \\ {H\left( {{jTs} + i + 1} \right)} \\ \vdots \\ {H\left( {{jTs} + i + L - 1} \right)} \end{bmatrix} \cdot {\underset{\_}{a}\left\lbrack {k - j} \right\rbrack}}} + {{\underset{\_}{b}}_{y}(t)}}} \end{matrix} \\ {{\underset{\_}{y}\left( {t = {{kTs} + i}} \right)} = {{\sum\limits_{j = 0}^{J - 1}{{H_{y}\left( {{jTs} + i} \right)} \cdot {\underset{\_}{a}\left\lbrack {k - j} \right\rbrack}}} + {{\underset{\_}{b}}_{y}(t)}}} \end{matrix}$

[0075] The estimator-demodulator 33 estimates the symbols ak] and detects their modulation states ã[k] and deduces by demodulation ^(Λ)v[k]. Device 41 of the space-time decoder 4 removes the apprenticeship sequences app. Device 42 then decodes the estimated useful symbols. Multiplexer 43 converts the decoded symbol vectors of dimension P into a flow of estimated data ^(Λ)d[i]. The position in the reception system of device 41 removing the apprenticeship sequences depends on the position in the transmission system 1 of device 14 adding these apprenticeship sequences.

[0076] The estimator-demodulator 33 can be made from traditional devices adapted to the model of the above expression of the space-time observation y(t), i.e. bi-dimensional. Two examples of realization are given on FIGS. 5a and 5 b.

[0077]FIG. 5a shows an estimator-demodulator 33 of symbols a[k] to a[k−J+1] in the sense of the least squares using the observation y(t): MMSE algorithm. The Wiener W filter 331 which satisfies a[k]=W.y(t) is first estimated by the filter coefficient estimation device 334 using the apprenticeship sequences app, then secondly applied outside these apprenticeship sequences to the space-time observations v(t) to estimate the vectors symbols a[k−J], (0≦j<J−1), whose modulation state is then detected by detectors 332 and lastly demodulated by demodulator 333.

[0078]FIG. 5b shows an estimator-demodulator 33 using the signals ã[k−Q+1] . . . {overscore (a)}[k−J+1] previously estimated and demodulated to estimate in the sense of the least squares the last Q signals a[k] . . . a[k−Q]: decision feed-back equalization (DFE) algorithm. Initialization of the filtering can be done by the last (J−Q) symbols of the apprenticeship sequences app. Once the vectors a[k] to a[k−Q] have been estimated, their states are detected and the symbols ã[k] . . . ã[k−Q] are deduced. The algorithm can therefore be summarized as follows:

[0079] Filter estimation: estimation of H_(y) from app by device 334,

[0080] Filtering initialization: ãee [k−Q+1] . . . {overscore (a)}[k−J+1]=app,

[0081] Instant k: Formation of y_(Q)(t) using filter 331B and the adder ${{\underset{\_}{y}}_{Q}(t)} = {{{\underset{\_}{y}(t)} - {\sum\limits_{j = Q}^{J - 1}{{H_{y}({jTs})} \cdot {\underset{\_}{\overset{\sim}{a}}\left\lbrack {k - j} \right\rbrack}}}} = {H_{y}^{Q} \cdot {{\overset{\sim}{\underset{\_}{a}}}_{Q}\lbrack k\rbrack}}}$ ${{with}\quad {{\overset{\sim}{\underset{\_}{a}}}_{Q}\lbrack k\rbrack}} = \left\lbrack {{\overset{\sim}{\underset{\_}{a}}\lbrack k\rbrack}^{T}\quad \cdots \quad {\overset{\sim}{\underset{\_}{a}}\left\lbrack {k - Q - 1} \right\rbrack}^{T}} \right\rbrack^{T}$

[0082] Estimation of a _(Q)[k] by filter 331T

â[k]=(H _(y) ^(Q) ·H _(y) ^(Qt))⁻¹ ·H _(y) ^(Qt)·y _(Q)(t)

[0083] Detection of the modulation states of â _(Q)[k] by detector 332

[0084] Demodulation by demodulator 333=>{overscore (a)} _(p)[k].

[0085] The coefficients of the transverse (331T) and recursive (331B) filters, respectively Ŵ_(Q) and Ĥ_(y) ^(Q) can be estimated:

[0086] Ŵ_(Q) with the same zero-forcing method by using Ĥ_(y) ^(Q) $\begin{matrix} {{\hat{W}}_{Q} = {R_{{\underset{\_}{app}}_{y} \cdot y}R_{y \cdot y}^{- 1}}} & {and} & {{\hat{H}}_{y}^{Q} = {R_{y \cdot {\underset{\_}{app}}_{y}}R_{{\underset{\_}{app}}_{y} \cdot {\underset{\_}{app}}_{y}}^{- 1}}} \end{matrix}$

[0087] Ŵ_(Q) in the sense of maximum resemblance with the Wiener method according to the following equation: $\begin{matrix} {{\hat{W}}_{Q} = {R_{{\underset{\_}{app}}_{y} \cdot y}R_{y \cdot y}^{- 1}}} & {and} & {{\hat{H}}_{y}^{Q} = {R_{y \cdot {\underset{\_}{app}}_{y}}R_{{\underset{\_}{app}}_{y} \cdot {\underset{\_}{app}}_{y}}^{- 1}}} \end{matrix}$

[0088] with R_(y.y) self-correlation of observations y_(Q)(t) containing the apprenticeship sequence app, and R _(app) _(y) _(·app) _(y) intercorrelation of the observations y(t) containing the apprenticeship sequence app and the apprenticeship sequence app.

[0089] Ĥ_(y) ^(Q) is estimated from the matrix Ĥ_(y)=[Ĥ_(y) ^(Q) . . . ]. Given that the matrix H_(y) is estimated in the sense of the least square: $\begin{matrix} {{\hat{W}}_{Q} = {R_{{\underset{\_}{app}}_{y} \cdot y}R_{y \cdot y}^{- 1}}} & {and} & {{\hat{H}}_{y}^{Q} = {R_{y \cdot {\underset{\_}{app}}_{y}}R_{{\underset{\_}{app}}_{y} \cdot {\underset{\_}{app}}_{y}}^{- 1}}} \end{matrix}$

[0090] where R _(app) _(y) _(·app) _(y) self-correlation of the apprenticeship sequence app, and R _(app) _(y) _(·app) _(y) and R _(app) _(y) _(·app) _(y) intercorrelation of the observations y(t) containing the apprenticeship sequence app and the apprenticeship sequence app.

[0091] A third example of realization could be an estimator-demodulator 33 comprising an estimator using the Viterbi type algorithm seeking all possible states of the set {a[k] . . . a[k−J+1]} which minimizes the difference between y(t) and H_(y·) a[k] and a demodulator 333 by deducing ^(Λ)v[k].

[0092] These three examples of realization are not limiting, the estimator must simply be able to take into account the two spatial and time dimensions of the observation y(t). For example, this space-time estimator of device 33 can be realized by a Viterbi type space-time algorithm or two dimensional filtering techniques (transverse filtering, decision feed-back filter, echo cancellation, etc.) for which the filters are estimated by algorithms of type MMSE, SGLS, RLS, Viterbi, Viterbi with weighted inputs and/or outputs, etc.

[0093] The transmission-reception system using such estimator-demodulators 33 operates irrespective of the channel, with a network of transmission-reception antennas colocalized or not, modulation being linear or linearizable, with or without memory, if the P modulator-transmitters have time diversity.

[0094] By introducing time diversity, the number of reception antennas N can be greater than, equal to or less than the number of transmission antennas P, in particular if different carrier frequencies are used for each transmitting antenna.

[0095] This transmission-reception system can be used to transmit digital signals in non-colocalized networks. It can also be used to transmit digital signals in colocalized networks if the number P of transmission antennas {24 ¹ . . . 24 ^(P)} is less than or equal to the number pf paths M (M≧1) of a transmitted signal transmitted by these transmission antennas on the transmission channel (P≦M), but also if the number P of transmission antennas {24 ¹ . . . 24 ^(P)} is greater than or equal to the number pf paths M (P≧M).

[0096] This transmission-reception system can be used to choose the transmission either of digital signals of several users or digital signals at high speed for one user. It is quite suitable for all types of network using several transmission antennas where it is necessary to choose between low, medium or high speed transmission for, for example, telephony, radio-broadcasting, television, transmission of interactive digital data (Internet), etc. irrespective of the network used such as, for example, the radio network, satellite, etc., in a transmission environment generating or not multiple reflections. 

1. Digital signal transmission system comprising: a space-time encoder (1) receiving a flow of data to be transmitted d[i], encoding this data d[i] as symbol vectors m[k] of dimension P (P>1) and generating said symbol vectors m[k], and P modulator-transmitters {2 ^(p)}_((1≦p≦P)), each receiving one component of the symbol vector m[k] output from the space-time encoder (1), applying the constellation of a predetermined modulation to said symbol m_(p)[k], and converting the symbol obtained a_(p)[k] into a signal s_(p)(t) transmitted on said antenna (24 ^(p)) connected to said transmitter (2 ^(p)) wherein the transmitters are adapted to transmit signals s(t) with time diversity.
 2. Transmission system according to claim 1, wherein these P modulator-transmitters {2 ^(p)}: each produce a symbol a_(p)[k] in parallel at instant k, each form a filter of function h_(p)(t), such that the function h_(p)(t) of the transmitter (2 ^(p)) is different from those of the other transmitters {2 ^(q)}_((q≠p)):: h₁(t)≠h₂(t)≠ . . . ≠h_(P)[(t), each generate at their respective transmission antennas (24 ^(p)) the signal s_(p)[k] corresponding at least to the filtering by the function h_(p)(t) of the symbols a_(p)[k].
 3. Transmission system according to the previous claim, wherein the function h_(p)(t) has one or more of the following features: any wave form h identical or not to that of the function h_(q)(t) of the transmitter (2 ^(q), q≠p) and a delay τ_(p) delaying the transmission of the received symbol a_(p)[k] by said duration τ_(p), such that the function h_(p)(t)=h(t−τ_(p)) with τ₁≠=τ₂≠ . . . τ_(p) for all values of p (1≦p≦P), any wave form h identical or not to that of the function h_(q)(t) of the transmitter (2 ^(q)) (q≠p) and a frequency shift f_(p) such that the function of the filter h_(p)(t)=h·exp(j2πf_(p)t) with f₁≠f₂ ≠ . . . f_(p) for all values of p (1≦p≦P), a wave form h_(p) with h₁≠h₂≠ . . . h_(p) for all values of p (1≦p≦P), (form of type NRZ, Nyquist with roll-off α or α_(p), etc.)
 4. Transmission system according to one of the previous claims: wherein said space-time encoder (1) comprises at least a demultiplexer with P channels (11) generating a symbol vector m[k] and/or at least one or more of the following devices: an encoder (12 ^(p)) generating a symbol vector c[k], a device (13) used to add at least an apprenticeship sequence app known by the receiver to the useful symbol vectors m[k] or encoded symbol vectors c[k] in order to create the symbol vectors v[k], and/or wherein each modulator-transmifter (2 ^(p)) comprises one or more of the following devices: a linear or linearizable modulator, a modulator with or without time memory, a BPSK or GMSK modulator, a device applying the constellation of said predetermined modulation (21 ^(p)) to the received symbols v_(e,uns p)[k] generating the symbols a_(p)[k], a device used to add at least the p^(th) component of an apprenticeship sequence app known by the receiver to the symbols a_(p)[k], a filter (22 ^(p)) filtering the constellated symbols a_(p)[k], a filter (22 ^(p)) filtering the vector of oversampled symbols $\begin{matrix} {{{\underset{\_}{u}}_{p}^{K}\left( {t = {{kTs} + i}} \right)} = {\begin{bmatrix} {u_{p}(t)} \\ \vdots \\ {u_{p}\left( {t - K + 1} \right)} \end{bmatrix} = \begin{bmatrix} {\underset{\_}{0}}_{i} \\ {a_{p}\lbrack k\rbrack} \\ {\underset{\_}{0}}_{{Ts} - 1} \\ {a_{p}\left\lbrack {k - 1} \right\rbrack} \\ \vdots \end{bmatrix}}} & {{{for}\quad 0} \leq i < {Ts}} \end{matrix}$ ${{with}\quad {\underset{\_}{0}}_{i}} = {{\begin{bmatrix} 0 & \cdots & 0 \end{bmatrix}^{T}\quad {and}\quad {\dim \left( {\underset{\_}{0}}_{i} \right)}} = {i \times 1}}$

an element (23 ^(p)) to modulate the signal to be transmitted on the carrier frequency f₀.
 5. Digital signal transmission method comprising: a space-time encoding step comprising at least the encoding of the flow of data to be transmitted d[i] as symbol vectors m[k] of dimension P (P>1), and a modulation-transmission step comprising at least: application in parallel of the constellation of a predetermined modulation to the P symbols m[k], transmission in parallel of the P signals s(t) obtained from the constellated symbols a[k] from P spatially separate points, wherein the modulation-transmission step is adapted to transmit the signals s(t) with time diversity.
 6. Transmission method according to the previous claim, wherein the modulation-transmission step comprises, in addition, at least, on each channel p (1≦p≦P), filtering of symbols a_(p)[k] generating a signal s_(p)(t) such that the filtering of channel p is different from that carried out on the other P-1 parallel channels.
 7. Transmission method according to the previous claim, wherein said filtering carried out on channel p has one or more of the following features: any wave form h identical or not to that of the channel q, for all values of p and q (1≦q≠p≦P) and a delay of duration τ_(p) different from that of channel q, any wave form h identical or not to that of the channel q, for all values of p and q (1≦q≠p≦P) and a frequency shift f_(p) different from that of channel q, a form h_(p) different from that of channel q, for all values of p and q (1≦q≠p≦P).
 8. Estimator-demodulator receiving in parallel N signals y(t) formed from L samples, wherein these signals y(t) represent a space-time observation since each of the N spatial components comprises L samples.
 9. Estimator-demodulator receiving in parallel N signals e,uns y(t) formed from L samples resulting from the transmission of digital signals by a transmission system according to one of claims 1 to 4, wherein these signals y(t) represent a space-time observation since each of the N spatial components comprises L samples.
 10. Estimator-demodulator of signals transmitted by a transmitter according to claim 8 or 9, wherein it comprises at least the following devices: a two dimensional Wiener filter estimator (334), said estimated two dimensional filter (331) Ŵ_(Q) receiving the observations y(t) and the estimation of the coefficients of said filter Ŵ_(Q), a detector (332) of the modulation states of the estimated symbols â[k]=Ŵ_(Q) y(t) generating the symbols detected {overscore (a)}[k], then a demodulator (333) generating the symbols ^(Λ)v[k],
 11. Estimator-demodulator of the signals transmitted by a transmitter according to claim 8 or 9, wherein it comprises at least the following devices: a first recursive two dimensional filter (331B) Ĥ_(y) ^(Q) receiving said estimation of its coefficients and the symbols already detected {overscore (a)}(k−Q) . . . {overscore (a)}(k−J+1), an adder used to remove the vector which is the result of the first filter applied to the received observations y(t) and obtain v _(q)(t), a second transverse two dimensional filter (331T) Ŵ_(Q) receiving y _(Q)(t) and said estimation of its coefficients and generating the estimated received symbols â _(Q)[k], a detector (332) of the modulation states of the estimated symbols â[k]=Ŵ_(q) y(t) generating the detected symbols ã[k], then a demodulator (333) generating the symbols ^(Λ)v[k], an estimator (334) of the coefficients Ŵ_(Q) of the transverse filter (331T) and Ĥ_(y) ^(Q) of the recursive filter (331B).
 12. Estimator-demodulator of the signals transmitted by a transmitter according to claim 8 or 9, wherein it comprises at least a two dimensional estimator-demodulator of the symbols {a[k] . . . a[k−J+1]} using a Viterbi algorithm.
 13. Estimator-demodulator according to one of claims 10 to 12, wherein it has one or more of the following features: the filter estimator (334) is an estimator either in the sense of maximum resemblance, the sense of least squares, or using a Viterbi algorithm, the demodulator (333) corresponds to a modulation of one or more of the following types: linear, linearizable, with time memory, without time memory, BPSK, GMSK, at least some of the received signals v(t) represent an apprenticeship sequences app known by said estimator-demodulator allowing one or more of the following operations: estimation of said filter (331), estimation in the sense of the least squares of said recursive filter (331B) such that its coefficients Ĥ_(y) ^(Q) are the first Q columns of the matrix ${{\hat{H}}_{y}^{Q} = {R_{y \cdot {\underset{\_}{app}}_{y}}R_{{\underset{\_}{app}}_{y} \cdot {\underset{\_}{app}}_{y}}^{- 1}}},$

estimation with the zero-forcing method of said transverse filter (331T) such that its coefficients are ${{\hat{W}}_{Q} = {R_{{\underset{\_}{app}}_{y} \cdot y}R_{y \cdot y}^{- 1}}},$

estimation using the Wiener method in the sense of maximum resemblance of said filter (331T) such that its coefficients are ${{\hat{W}}_{Q} = {R_{{\underset{\_}{app}}_{y} \cdot y}R_{y \cdot y}^{- 1}}},$

initialization of the estimation of said filter(s) (331 or 331T and 331B), initialization of said filtering (331B), initialization of the Viterbi algorithm of the estimator-demodulator (33).
 14. Digital signal reception system comprising: a receiver (3) comprising at least a network of N reception antennas and an estimator-demodulator (33) according to one of claims 8 to 13, and a space-time decoder (4) wherein said receiver (3) comprises at least: N reception devices {31 ^(n)}_((1≦n≦N)), comprising at least an element used to put the received signal in the baseband, generating an observation vector x(t) of dimension N, a windower (32) producing from observations x(t) the discrete observations x[kTs+i] with t=kts+i and 0≦i ≦Ts given that the observations x[kTs+i] depend on the transmitted signal vectors a[k] to a[k−J+1] and generating a space-time observation ${\underset{\_}{y}(t)} = \begin{bmatrix} {\underset{\_}{x}(t)} \\ {\underset{\_}{x}\left( {t - 1} \right)} \\ \vdots \\ {\underset{\_}{x}\left( {t + L - 1} \right)} \end{bmatrix}$

from N observations x(t), and/or wherein the space-time decoder (4) comprises at least one or more of the following devices: an element (41) capable of removing the apprenticeship sequences app, a decoder (42) with P channels in input/output, a multiplexer (43) with P channels in input.
 15. Estimation and demodulation method comprising a step of reception in parallel of N signals y(t), wherein the observation y(t) is space-time since each of the N spatial components comprises L samples.
 16. Estimation and demodulation method comprising a step of reception in parallel of N signals y(t) resulting from the transmission according to the transmission method of one of claims 5 to 7, wherein the observation y(t) is space-time since each of the N spatial components comprises L samples.
 17. Estimation and demodulation method according to claim 15 or 16, wherein it uses one of the following two dimensional algorithms: a direct estimation algorithm of the a[k] in the sense of the least squares from the observation y(t), a decision feed-back algorithm, i.e. using demodulated symbol vectors for estimation in the sense of the least squares, a Viterbi algorithm.
 18. Estimation and demodulation method according to claim 17, wherein: the demodulation carried out on the estimated symbols corresponds to a linear or linearizable modulation with or without time memory or BPSK or GMSK, and/or wherein at least some of the received signals represent an apprenticeship sequence used by said algorithms.
 19. Digital signal transmission system comprising at least a transmission system according to one of claims 1 to 4, and a reception system either comprising an estimator-demodulator according to one of claims 8 to 13, or according to claim 14, wherein it comprises, in addition, at least a transmission channel such that a signal s_(p)(t) transmitted by said transmission system takes M separate paths (M≧1) in said transmission channel before reaching said reception system.
 20. Transmission system according to the previous claim, wherein the number of transmission antennas P of said transmission system is greater than or equal to the number of paths M (P≧M).
 21. Transmission system according to the previous claim, wherein the number of transmission antennas P of said transmission system is less than or equal to the number of paths M (P≦M).
 22. Use of the transmission system according to one of claims 1 to 4 and of the reception system either comprising an estimator-demodulator according to one of claims 8 to 13, or according to claim 14 to the transmission of digital signals on a transmission channel such that a signal s_(p)(t) transmitted by said transmission system takes M separate paths (M≧1) in said transmission channel before reaching said reception system and such that the number of transmission antennas P of said transmission system is greater than or equal to the number of paths M (P≧M).
 23. Use of the transmission system according to one of claims 1 to 4 and of the reception system either comprising an estimator-demodulator according to one of claims 8 to 13, or according to claim 14 to the transmission of digital signals on a transmission channel such that a signal s_(p)(t) transmitted by said transmission system takes M separate paths (M≧1) in said transmission channel before reaching said reception system and such that the number of transmission antennas P of said transmission system is greater than or equal to the number of paths M (P≦M). 